The usage of light-emitting diodes (LEDs) to provide illumination is increasing rapidly as the cost of LEDs decrease and the endurance of the LEDs increases to cause the overall effective cost of operating LED lighting products to be lower than incandescent lamps and fluorescent lamps providing equivalent illumination. Also, LEDs can be dimmed by controlling the current through the LEDs because LEDs are current driven devices. The current through a plurality of LEDs in a lighting device must be controlled tightly in order to control the illumination provided by the LEDs. Typically, the secondary of an LED lighting device must be electrically isolated from the primary (line and neutral side) of the lighting device to meet applicable safety standards (e.g., IEC class II isolation). In addition, an LED driver circuit should have a high power factor and should have a constant current control.
One known solution to the foregoing requirements is to use a flyback converter to produce the DC in the secondary from the primary source. The flyback converter provides power factor correction to produce a high power factor, provides isolation between the primary and secondary circuits, and has a reasonably low cost. The flyback converter uses secondary current sensing techniques with feedback to the primary converter to control the secondary current through the LEDs. The flyback converter provides an LED driver that is low in cost when compared with other topologies.
FIG. 1 illustrates a traditional LED driver circuit 100 based on a flyback converter. The LED driver circuit includes a primary section 102 and a secondary section 104. The LED driver circuit provides current to an LED load 110. In the illustrated embodiment, the LED load may comprise from two to five LEDs 112 connected in series between a first (+) LED load terminal 114 and a second (−) LED load terminal 116. A common load current flows through each LED in the LED load to cause the LEDs to illuminate. In alternative embodiments, the LED load may comprise additional LEDs in series or a series-parallel combination of LEDs. In order to provide consistent illumination, the load current through the LEDs should be maintained at a substantially constant magnitude. The illustrated driver circuit utilizes a secondary current sensing technique (described below) to control the secondary current.
In the LED driver circuit 100, an AC source 120 provides an AC input voltage via a first AC input line 122 and a second AC input line 124. In the illustrated embodiment, the AC input voltage may vary from 86 volts RMS to 265 volts RMS. The AC input voltage between the first AC input line and the second AC input line is applied between a first input terminal 132 and a second input terminal 134 of a full-wave bridge rectifier 130. The bridge rectifier has a first (+) output terminal 136 and a second (−) output terminal 138. A first rectifier diode 140 has an anode connected to the first input terminal and a cathode connected to the first output terminal. A second rectifier diode 142 has an anode connected to the second input terminal and a cathode connected to the first output terminal. A third rectifier diode 144 has an anode connected to the second output terminal and has a cathode connected to the first input terminal. A fourth rectifier diode 146 has an anode connected to the second output terminal and has a cathode connected to the second input terminal. The bridge rectifier operates in a conventional manner to produce a pulsating DC voltage on the first output terminal which is referenced to the second output terminal. The second output terminal is connected to a primary ground reference 150.
The first (+) output terminal 136 of the bridge rectifier 130 is connected to a first terminal 164 of the primary winding 162 of an isolation transformer 160, which galvanically isolates the primary section 102 of the LED driver circuit 100 from the secondary section 104. The primary winding of the isolation transformer has a second terminal 166. The isolation transformer has a secondary winding 170, which has a first terminal 172 and a second terminal 174. The isolation transformer has an N:1 turns ratio between the primary winding and the secondary winding such that the voltage across the primary winding is N times the voltage across the secondary winding and such that the current through the secondary winding is N times the current through the primary winding.
As further illustrated in FIG. 1, the first terminal 172 of the secondary winding 170 of the isolation transformer 160 is connected to secondary ground reference 180. The secondary ground reference is electrically isolated from the primary ground reference 150 by the isolation transformer. The second terminal 174 of the secondary winding is connected to the anode of a secondary diode 182. The cathode of the secondary diode is connected to the first (+) terminal 186 of a secondary filter capacitor 184. The secondary filter capacitor may also be referred to as an output filter capacitor. A second (−) terminal 188 of the secondary filter capacitor is connected to the secondary ground reference and thus to the first terminal of the secondary winding of the isolation transformer. In one embodiment, the secondary filter capacitor has a capacitance of approximately 2,000 microfarads. The cathode of the secondary diode and the first terminal of the secondary filter capacitor are connected to a first (+) output terminal 190 of the LED driver circuit 100. The secondary ground reference is connected to a second (−) output terminal 192 of the LED driver circuit via a current sensing resistor 200.
As illustrated in FIG. 1, dots on the terminals of the primary winding 162 and the secondary winding 170 of the isolation transformer represent the magnetic coupling between the two windings. When the first terminal 164 of the primary winding is positive with respect to the second terminal 166 of the primary winding, the first terminal 172 of the secondary winding is also positive with respect to the second terminal 174 of the secondary winding; however, the current flow through the secondary winding is opposite the current flow through the primary winding. Thus, when current flows into the first terminal of the primary winding and flows to the second terminal of the primary winding with an increasing magnitude, the increasing current flow should induce current to flow from the second terminal of the secondary winding to the first terminal of the secondary winding (e.g., downward through the secondary winding toward the secondary ground reference 180) when the secondary winding terminals are oriented as shown in FIG. 1). However, induced current flow in that direction is blocked by the reverse-biased secondary diode 182. In contrast, when the magnitude of the current flowing from the first terminal to the second terminal of the primary winding decreases, current flow is induced in the secondary winding that flows from the first terminal of the secondary winding to the second terminal of the secondary winding (e.g., upward through the secondary winding when the secondary winding terminals are oriented as shown in FIG. 1). The current flowing out of the second terminal of the secondary winding passes through the forward-biased secondary diode and charges the secondary filter capacitor 184.
The first (+) output terminal 190 of the LED driver circuit 100 is connected to the first (+) terminal 114 of the LED load 110. The second (−) terminal 116 of the LED load is connected to the second (−) output terminal 192 of the LED driver circuit. Thus, the second terminal of the LED load is connected to the secondary ground reference 180 via the current sensing resistor 200. Accordingly, when an output current IOUT flows through the LED load, a voltage (VISENSE) develops across the current sensing resistor with respect to the secondary ground reference. The voltage across the current sensing resistor appears on the second (−) output terminal of the LED driver circuit. The sensed voltage is proportional to the magnitude of the current flowing through the LED load. In the illustrated LED driver circuit, the current sensing resistor is very small (e.g., approximately 0.1 ohm to approximately 3 ohms) such that the power loss in the resistor is very small (e.g., less than about 100 milliwatts) and such that the voltage drop across the current sensing resistor is also very small and does not substantially affect the voltage available to the LED load. For example, for a resistance of 0.1 ohm and a load current of approximately 180 milliamperes, the sensed voltage drop across the current sensing resistor is only approximately 18 millivolts. The current sensing resistor generates a voltage with respect to the secondary ground reference that is proportional to the current flowing through the sensing resistor.
It should be understood that the current sensing resistor 200 may be located at other positions in the current path from the filter capacitor 184 to the LED load 110 and back to the filter capacitor. The illustrated location with the sensing resistor in the current return path allows the voltage to be measured with respect to the secondary ground reference 180 using a single-ended amplifier (described below). Alternatively, the sensing resistor may be located, for example, between the first terminal of the filter capacitor and the first output terminal. In the absence of the connection to the secondary ground reference, the voltage across the sensing resistor may be measured by a differential amplifier.
The second (−) terminal 116 of the LED load 110 is also connected to an input terminal 212 of a sensing voltage amplifier (AMP) 210. In the illustrated embodiment, the sensing voltage amplifier may comprise a first conventional operational amplifier (OpAmp), which is configured as a single-ended amplifier to buffer and amplify the relatively small voltage developed across the current sensing resistor 200 with respect to the secondary ground reference 180 to provide an output voltage on an output terminal 214. In an alternative embodiment (not shown), the sensing voltage amplifier may be configured as a differential amplifier to sense the voltage across the sensing resistor without reference to the secondary ground reference.
The output terminal 214 of the sensing voltage amplifier 210 is connected to a first input terminal 222 of a voltage difference circuit (DIF) 220, which has a second input terminal 224 and an output terminal 226. The second input terminal of the voltage difference circuit is connected to an output terminal 232 of a reference source (REF) 230. In the illustrated embodiment, the voltage difference circuit may comprise a second conventional OpAmp, which is configured to output a voltage on the output terminal that is responsive to a difference between voltages on the first and second input terminals. The reference source provides a reference output voltage VIREF that is proportional to a desired current through the LED load. For example, in one embodiment of the driver circuit 100, the desired current through the LED load is 180 milliamps. The reference voltage corresponding to the desired current is selected such that when 180 milliamps flows through the current sensing resistor 200, the sensing voltage (VISENSE) developed across the current sensing resistor as amplified by the sensing voltage amplifier is substantially equal to the reference voltage (VIREF). The voltage difference circuit outputs a first (nominal) voltage when the two input voltages are substantially equal. If the amplified sensing voltage differs from the reference voltage, the voltage difference circuit outputs a voltage having an amplitude that differs from the first voltage by a magnitude and a direction (e.g., more than or less than the first voltage) responsive to the difference between the two input voltages. For example, in one embodiment, the first (nominal) voltage may be set to approximately one-half of the rail-to-rail supply voltage of the second OpAmp such that the output of the second OpAmp is a voltage that varies with respect to the nominal voltage.
The output voltage on the output terminal 226 of the voltage difference circuit 220 is provided as an input to an optical isolator (e.g., an optical coupler) 240. For example, in the illustrated embodiment, the output of the voltage difference circuit is connected to the anode of an input LED 242 of the optical isolator. The cathode of the input LED of the optical isolator is connected to the secondary ground reference 180 via a current limiting resistor 244. The optical isolator has an optically isolated output transistor 250 that controls a collector-emitter voltage in response to the magnitude of the current flowing through the input LED. The emitter of the output transistor is connected to the primary ground 150 via an emitter biasing resistor 252. The collector of the output transistor is connected to a first terminal of a pullup resistor 262 at a primary feedback node 260. A second terminal of the pullup resistor is connected to a logic circuit supply voltage (VCC). The operation of the optical isolator causes the voltage at the collector of the output transistor—and thus the voltage at the primary feedback node—to vary in response to the output voltage of the voltage difference circuit. Thus, the voltage at the primary feedback node has a magnitude that represents the difference between the two input voltages on the inputs of the voltage difference circuit.
As further shown in FIG. 1, the second terminal 166 of the primary winding 162 of the isolation transformer 160 is connected to a first terminal 302 of a semiconductor switch 300. The switch further includes a second terminal 304 and a control terminal 306. For example, the semiconductor switch may comprise a metal oxide semiconductor field effect transistor (MOSFET) wherein the first terminal is the drain of the MOSFET, the second terminal is the source of the MOSFET, and the control terminal is the gate of the MOSFET. In the illustrated embodiment, the MOSFET is an N-channel enhancement mode transistor, which has is normally off (e.g., has a high impedance between the drain and the source). The MOSFET turns on to provide a low-impedance path (e.g., a few tens of milliohms) between the drain and the source when a sufficiently large voltage differential is applied between the gate and the source of the MOSFET. The second terminal (source) of the MOSFET is connected to the primary ground reference 150. When the MOSFET is turned on, a current flows from the first (+) output terminal 136 of the bridge rectifier 130, through the primary winding 162 of the isolation transformer 160, through the MOSFET from the first terminal (drain) to the second terminal (source), and to the primary ground reference.
The control terminal (gate) 306 of the MOSFET 300 is controlled by a gate drive (GD) output terminal 322 of a switch controller integrated circuit (“switch controller IC”) 320. In the illustrated embodiment, the switch controller IC comprises an L6562 transition-mode power factor correction (PFC) controller, which is commercially available from STMicroelectronics of Geneva, Switzerland. The switch controller IC receives a feedback voltage via a feedback input terminal (INV) 324, which is connected to the primary feedback node 260. Thus, the switch controller IC receives a voltage responsive to the difference between the instantaneous LED load current flowing through the current sensing resistor 200 and the desired LED load current. The switch controller IC further includes a zero-crossing detector (ZCD) input terminal 330 that is connected to a first terminal 334 of a third (auxiliary) winding 332 of the isolation transformer 160 via a resistor 338. A second terminal 336 of the third winding is connected to the primary ground reference 150. The auxiliary winding produces a voltage responsive to the voltage on the primary winding 164 of the isolation transformer. Circuitry within the switch controller IC 320 detects when the voltage on the auxiliary winding is at zero volts, and modifies the internal switching operations to control the power factor of the line current in a known manner. Information regarding the power factor control function of the switch controller IC is provided by the manufacturer. The power factor control function is not pertinent to the present disclosure. The switch controller IC includes additional inputs (e.g., power input, ground reference, and compensation inputs), which are not shown in FIG. 1.
The switch controller IC 320 operates in a conventional manner to output a high output signal on the gate control output terminal 322 to turn on the MOSFET 300 to cause current to flow through the primary winding 162 of the isolation transformer 160 from the first terminal 164 to the second terminal 166 of the primary winding. The switch controller IC output a low output signal on the gate control output terminal to turn off the MOSFET to stop current flow through the primary winding of the transformer. The time varying current flow through the primary winding generates current flow in the secondary winding 170, which is rectified by the secondary diode 182 and which is applied to the secondary filter capacitor 184 to thereby charge the secondary filter capacitor. The voltage across the secondary filter capacitor is applied to the LED load 110 to cause the output current IOUT to flow through the load.
The output current IOUT flowing through the LED load 110 is sensed by the current sensing resistor 200. The voltage VISENSE representing the sensed current is amplified and compared to a reference signal to produce a feedback signal, which is applied to the feedback input (INV) of the switch controller IC, as described above. The switch controller IC is responsive to the feedback signal to switch the MOSFET on and off with varying durations to adjust the voltage across the secondary filter capacitor to a magnitude sufficient to cause the current flowing through the LED load to have a desired magnitude (e.g., 180 milliamps in the illustrated example). Note that although the operation of the switch controller IC determines the voltage across the secondary filter capacitor, the actual voltage across the LED load required to maintain the desired current through the LED load varies with the characteristics of the LEDs within the LED load and also varies with other factors such as, for example, temperature. Thus, it should be understood that the sensed output current through the LED load is the controlled parameter. The secondary voltage across the LED load may vary to maintain the sensed current magnitude at or near the desired output current magnitude (e.g., at approximately 180 milliamperes in certain embodiments).
The LED driver circuit 100 shown in FIG. 1 uses the flyback converter with secondary current sensing to provide a simple and cost effective way to control the current through the LED load 110; however, testing has shown that the circuit in FIG. 1 does not provide adequate current regulation over a wide range of input voltages and over a range of the number of LEDs 112 in series within in the LED load. Furthermore, the ripple in the output current through the LED load may be excessive under certain load condition. For example, at lighter loads (e.g., fewer LEDs in the LED load 100), the ripple percentage increases and the range of output currents applied to the load increases.
The foregoing is illustrated in Table 1, which shows the effects on the output current caused by changing the number of LEDs connected in series within the LED load 110. In the following table, the secondary filter capacitor 184 has a capacitance of approximately 2,000 microfarads, and each LED has a nominal forward voltage of between 2.75 volts and 2.8 volts.
TABLE 1No. ofIOUT_AVGIOUT_MAXIOUT_MINVOUTRippleLEDs(mA)(mA)(mA)(volts)(%)518020915013.8616.11418022014911.122.2231802351458.430.5621802551435.641.67
As illustrated in Table 1, the target output current through the LED load 100 is 180 milliamps. The input voltages from the AC source 120 range from approximately 86 volts RMS to approximately 265 volts RMS. In the illustrated embodiment, the LED load 110 may include as few as two LEDs 112 in series (requiring approximately 5.6 volts across the LED load) to as many as five LEDs in series (requiring approximately 13.9 volts across the LED load). The testing data in Table 1 show that when the LED load includes 5 LEDs in series, the LED driver circuit 100 is able to maintain the output current through the LED load within a range from 150 milliamps to 209 milliamps—a range from 30 milliamps less than the nominal current to 29 milliamps greater than the nominal current. With 5 LEDs in the LED load, the ripple in the output current is approximately 16.11%.
Decreasing the number of LEDs in the LED load 110 decreases the voltage across the LED load. At lower load voltages, the flyback converter in the LED driver 100 is unable to regulate the output current and control the ripple in the output current as well as at the higher load voltage. For example, with only 4 LEDs in series having an overall load voltage of approximately 11.1 volts, the output current varies over a greater range of magnitude from approximately 149 milliamps to approximately 220 milliamps; and the output current has a ripple of approximately 22.22%. With only 2 LEDs in series having an overall load voltage of approximately 5.6 volts, the output current varies over an even greater range of magnitudes from approximately 143 milliamps to approximately 255 milliamps; and the output current has a ripple of approximately 41.67%.
The maximum current of 255 milliamps in Table 1 exceeds a typical peak current rating of 240 milliamps for an LED having a nominal current rating of 180 milliamps. Furthermore, the ripple of the load current for each system with fewer than 5 LEDs in series exceeds a typical requirement that the ripple on the current through the LEDs be less than 20%. Accordingly, a need exists to be able to operate the flyback converter in the LED driver 100 with as few as 2 LEDs in series while maintaining the maximum current through the LEDs at or below 240 milliamps, and also to operate the flyback converter with as few as 2 LEDs in series while maintaining the ripple on the current to less than 20%.
One simple way of reducing the ripple is to increase the capacitance of the secondary filter capacitor 184. For example, the capacitance can be increased to 4,000 microfarads. Although this solution may be simple in concept, the solution is costly economically and increases circuit board space, increases component height or increases both circuit board space and component height. Electrolytic capacitors with greater capacitance typically cost more and occupy more volume. The increased volume may require a larger diameter on a printed circuit board or may require a greater height above a printed circuit board or may require both a greater board area and a greater height. Accordingly, a solution is desired that reduces the current ripple and improves the current regulation without increasing the capacitance of the secondary filter capacitor.